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6-bit, programmable 2-/3-/4-phase, synchronous buck controller ADP3194 features selectable 2-, 3-, or 4-phase operation up to 1 mhz per phase 9.5 mv worst-case differential sensing error over temperature logic-level pwm outputs for interface to external high power drivers pwm flex-mode tm architecture for excellent load transient performance active current balancing between all output phases built-in power good/crowbar blanking supports otf vid code changes 6-bit digitally programmable 0.8375 v to 1.6 v output programmable short circuit protection with programmable latch-off delay applications desktop pc power supplies for next-generation intel? processors vrm modules games consoles functional block diagram v cc gnd ADP3194 en delay ilimit p wrg d rt rampadj pwm2 fb pwm3 pwm4 sw1 cssum cscomp sw2 sw3 sw4 csref pwm1 vid4 vid3 vid2 vid1 vid5 vid0 fbrtn comp dac + 150mv dac ? 250mv csref en crowbar current limit reset reset reset reset 2-/3-/4-phase driver logic en set curren t- balancing circuit oscillator delay uvlo shutdown and bias curren t- limit circuit soft start precision reference vid dac cmp cmp cmp cmp 28 13 14 26 8 25 24 23 17 18 22 21 20 16 27 1 2 3 4 6 5 7 19 11 12 15 10 9 06022-001 shunt regulator figure 1. functional block diagram general description the ADP3194 1 is a highly efficient, multiphase, synchronous buck switching regulator controller optimized for converting a 5 v or 12 v main supply into the core supply voltage required by high performance intel processors. it uses an internal 6-bit dac to read a voltage identification (vid) code directly from the processor that is used to set the output voltage between 0.8375 v and 1.6 v. the device uses a multimode pwm archi- tecture to drive the logic-level outputs at a programmable switching frequency that can be optimized for vr size and efficiency. the phase relationship of the output signals can be programmed to provide 2-, 3-, or 4-phase operation, allowing for the construction of up to four complementary buck switch- ing stages. the ADP3194 also includes programmable, no-load offset, and slope functions to adjust the output voltage as a function of the load current, so it is always optimally positioned for a system transient. the ADP3194 also provides accurate and reliable short-circuit protection, adjustable current limiting, and a delayed power good output that accommodates on-the- fly (otf) output voltage changes requested by the cpu. the devices are specified over the commercial temperature range of 0c to +85c and are available in a 28-lead tssop. 1 protected by u. s. patent numb er 6,683,441; other patents pending. rev. 0 information furnished by analog devices is believed to be accurate and reliable. however, no responsibility is assumed by analog devices for its use, nor for any infringements of patents or other rights of third parties that may result from its use. specifications subject to change without notice. no license is granted by implication or otherwise under any patent or patent rights of analog devices. trademarks and registered trademarks are the property of their respective owners. one technology way, p.o. box 9106, norwood, ma 02062-9106, u.s.a. tel: 781.329.4700 www.analog.com fax: 781.461.3113 ?2006 analog devices, inc. all rights reserved.
ADP3194 rev. 0 | page 2 of 32 table of contents features .............................................................................................. 1 applications....................................................................................... 1 functional block diagram .............................................................. 1 general description ......................................................................... 1 revision history ............................................................................... 2 specifications..................................................................................... 3 absolute maximum ratings............................................................ 5 esd caution.................................................................................. 5 pin configuration and function descriptions............................. 6 typical performance characteristic and test circuits ................ 7 theory of operation ........................................................................ 8 startup sequence .......................................................................... 8 master clock frequency.............................................................. 8 output voltage differential sensing .......................................... 8 output current sensing .............................................................. 8 active impedance control mode............................................... 9 current-control mode and thermal balance.......................... 9 voltage control mode.................................................................. 9 soft start ........................................................................................ 9 current-limit, short-circuit, and latch-off protection...... 10 dynamic vid.............................................................................. 10 power good monitoring ........................................................... 12 output crowbar ......................................................................... 12 output enable and uvlo ........................................................ 12 application information................................................................ 13 setting the clock frequency ..................................................... 13 soft start and current-limit latch-off delay times ........... 13 inductor selection ...................................................................... 13 designing an inductor............................................................... 14 sense resistor.............................................................................. 14 sense resistor selection ............................................................ 14 output droop resistanceCsense resistor............................... 14 output offset .............................................................................. 15 design comparison trade-off between dcr and sense resistor ........................................................................................ 15 c out selection ............................................................................. 16 ramp resistor selection............................................................ 17 comp pin ramp ....................................................................... 17 current-limit setpoint.............................................................. 17 feedback loop compensation design.................................... 17 c in selection and input current di/dt reduction.................. 19 tuning the ADP3194 ................................................................. 20 rampadj filter......................................................................... 22 shunt resistor design................................................................ 22 design example using dcr method.......................................... 23 inductor selection using dcr................................................. 23 designing an inductor using dcr.......................................... 23 inductor dcr temperature correction ................................. 23 output droop resistanceCdcr method................................ 24 power mosfets......................................................................... 24 layout and component placement.............................................. 26 general recommendations....................................................... 26 power circuitry recommendations ........................................ 26 signal circuitry recommendations......................................... 26 outline dimensions ....................................................................... 29 ordering guide .......................................................................... 29 revision history 10/06revision 0: initial version
ADP3194 rev. 0 | page 3 of 32 specifications vcc = 5 v, fbrtn = gnd, t a = 0c to +85c, unless otherwise noted. 1 table 1. parameter symbol conditions min typ max unit error amplifier output voltage range v comp 0 vcc v accuracy v fb relative to nominal dac output, referenced to fbrtn, cssum = cscomp; v out < 1 v ?8.0 +8.0 mv accuracy v fb relative to nominal dac output, referenced to fbrtn, cssum = cscomp; v out > 1 v ?9.5 +9.5 mv line regulation v fb vcc = 4.75 v to 5.25 v 0.05 % input bias current i fb 14 15.5 17 a fbrtn current i fbrtn 100 140 a output current i o(err) fb forced to v out C 3% 500 a gain bandwidth product gbw (err) comp = fb 20 mhz slew rate c comp = 10 pf 25 v/s vid inputs input low voltage v il(vid) 0.4 v input high voltage v ih(vid) 0.8 v input current, input voltage low i il(vidx) vid(x) = 0 v C25 C35 a input current, input voltage high i ih(vidx) vid(x) = 1.25 v 5 15 a pull-up resistance r vid 35 60 85 k internal pull-up voltage 1.0 1.2 v vid transition delay time 2 vid code change to fb change 400 ns no cpu detection turn-off delay time 2 vid code change to 11111 to pwm going low 400 ns oscillator frequency range 2 f osc 0.25 4.5 mhz frequency variation f phase t a = +25c, r t = 247 k, 4-phase 1.55 2 2.45 mhz t a = +25c, r t = 138 k, 4-phase 3 mhz t a = +25c, r t = 84 k, 4-phase 4 mhz output voltage v rt r t = 100 k to gnd 1.8 2.0 2.3 v rampadj output voltage v rampadj rampadj C fb C50 +50 mv rampadj input current range i rampadj 0 100 a current sense amplifier offset voltage v os(csa) cssum C csref C1.5 +1.5 mv input bias current i bias(cssum) C10 +10 na gain bandwidth product gbw (csa) 10 mhz slew rate c cscomp = 10 pf 10 v/s input common-mode range cssum and csref 0 3 v positioning accuracy v fb C77 C80 C83 mv output voltage range 0.05 vcc v output current i cscomp 500 a current balance circuit common-mode range v sw(x)cm C600 +200 mv input resistance r sw(x) sw(x) = 0 v 12 20 28 k input current i sw(x) sw(x) = 0 v 5 11 17 a input current matching 3 i sw(x) sw(x) = 0 v C5 +5 %
ADP3194 rev. 0 | page 4 of 32 parameter symbol conditions min typ max unit current limit comparator output voltage normal mode v ilimit(nm) en > 0.8 v, r ilimit = 250 k 2.8 3 3.3 v shutdown mode v ilimit(sd) en < 0.4 v, i ilimit = C100 a 400 mv output current, normal mode i ilimit(nm) en > 0.8 v, r ilimit = 250 k 12 a maximum output current 2 60 a current limit threshold voltage v cl v csref C v cscomp , r ilimit = 250 k 105 125 145 mv current limit setting ratio v cl /i ilimit 10.4 mv/a delay normal mode voltage v delay(nm) r delay = 250 k 2.8 3 3.3 v delay overcurrent threshold v delay(oc) r delay = 250 k 1.6 1.9 2.2 v latch-off delay time t delay r delay = 250 k, c delay = 12 nf 1.5 ms soft start output current, soft start mode i delay(ss) during startup, delay < 2.8 v 15 20 25 a soft start delay time t delay(ss) r delay = 250 k, c delay = 12 nf, vid code = 011111 1 ms enable input input low voltage v il(en) 0.4 v input high voltage v ih(en) 0.8 v input current i il(en) C1 +1 a power good comparator undervoltage threshold v pwrgd(uv) relative to nominal dac output C180 C250 C300 mv overvoltage threshold v pwrgd(ov) relative to nominal dac output 90 150 200 mv output low voltage v ol(pwrgd) i pwrgd(sink) = 4 ma 225 400 mv power good delay time during soft start r delay = 250 k, c delay = 12 nf, vid code = 011111 1 ms vid code changing 100 250 s vid code static 200 ns crowbar trip point v crowbar relative to nominal dac output 90 150 200 mv crowbar reset point relative to fbrtn 450 550 650 mv crowbar delay time t crowbar overvoltage to pwm going low vid code changing blanking time 100 250 s vid code static 400 ns pwm outputs output low voltage v ol(pwm) i pwm(sink) = C400 a 160 500 mv output high voltage v oh(pwm) i pwm(source) = +400 a 4.0 5 v supplyADP3194 v system = 12 v, r shunt = 300 vcc vcc 4.75 5 v dc supply current 20 30 ma uvlo threshold voltage v uvlo vcc rising 6.3 7 8.0 v uvlo hysteresis 0.9 v 1 all limits at temperature extremes ar e guaranteed via correlation using standard statistical quality control (sqc). 2 guaranteed by design, not production tested. 3 relative current matching from each phase to the average of all four phases.
ADP3194 rev. 0 | page 5 of 32 absolute maximum ratings table 2. parameter rating vcc C0.3 v to +6 v vid4 to vid0, vid5 C0.3 v to +6 v fbrtn C0.3 v to +0.3 v sw1 to sw4 ?5 v to +25 v all other inputs and outputs C0.3 v to vcc + 0.3 v storage temperature range C65c to +150c operating ambient temperature range 0c to +85c operating junction temperature 125c thermal impedance ( ja ) 100c/w lead temperature soldering (10 sec) 300c vapor phase (60 sec) 215c infrared (15 sec) 220c stresses above those listed under absolute maximum ratings may cause permanent damage to the device. this is a stress rating only; functional operation of the device at these or any other conditions above those indicated in the operational section of this specification is not implied. exposure to absolute maximum rating conditions for extended periods may affect device reliability. absolute maximum ratings apply individually only, not in combination. unless otherwise specified, all other voltages are referenced to gnd. esd caution
ADP3194 rev. 0 | page 6 of 32 pin configuration and fu nction descriptions ADP3194 top view (not to scale) 1 2 3 28 27 26 4 8 10 12 14 5 6 7 21 24 23 22 9 11 17 18 19 13 15 16 20 25 vid4 vid3 vid2 vid1 vid0 vid5 fbrtn fb comp pwrgd en delay rt rampadj vcc pwm1 pwm2 pwm3 pwm4 sw1 sw2 sw3 sw4 gnd cscomp cssum csref ilimit 0 6022-002 figure 2. pin configuration table 3. pin function descriptions pin no. mnemonic description 1 to 6 vid4 to vid0, vid5 voltage identification dac inputs. these six pins are pull ed up to an internal reference, providing logic 1 is left open. when in normal opera tion mode, the dac output programs the fb regulation voltage from 0.8375 v to 1.6 v (see table 2 ). leaving all the vid pins open results in ADP3194 going into a no cpu mode, shutting off their pwm outputs and pulling the pwrgd output low. 7 fbrtn feedback return. vid dac and error amplifier reference for remote sensing of the output voltage. 8 fb feedback input. error amplifier input for remote sensin g of the output voltage. an external resistor between this pin and the output voltage sets the no-load offset point. 9 comp error amplifier outp ut and compensation point. 10 pwrgd power good output. open drain output that signals when the output voltage is outside of the proper oper- ating range. 11 en power supply enable input. pulling this pin to gnd disables the pwm outputs and pulls the pwrgd output low. 12 delay soft start delay and current-limit latc h-off delay setting input. an extern al resistor and capacitor connected between this pin and gnd sets the soft start ramp-up time and the overcurrent latch-off delay time. 13 rt frequency setting resistor input. an external resistor connected between this pin and gnd sets the oscillator frequency of the device. 14 rampadj pwm ramp current input. an external resistor from the converter input voltage to this pin sets the internal pwm ramp. 15 ilimit current limit set point/enable output. an external resist or from this pin to gnd sets the current limit thresh old of the converter. this pin is actively pulled low when the ADP3194 en input is low or when vcc is below its uvlo threshold to signal to the driver ic that th e driver high-side and low-side outputs should go low. 16 csref current sense reference voltage input. the voltage on this pin is used as the reference for the current sense amplifier and the power good and crowbar functions. this pin should be connected to the common point of the output inductors. 17 cssum current sense summing node. external resistors from each switch node to this pin sum the average inductor currents together to measure the total output current. 18 cscomp current sense compensation point. a resistor and capacito r from this pin to cssum determines the slope of the load line and the positioning loop response time. 19 gnd ground. all internal biasing and the logic output si gnals of the device are referenced to this ground. 20 to 23 sw4 to sw1 current balance inputs. inputs for me asuring the current level in each ph ase. leave the sw pins of unused phases open. 24 to 27 pwm4 to pmw1 logic level pwm outputs. each output is connected to th e input of an external mosfet driver such as the adp3120a. connecting the pwm3 and/or pwm4 outputs to gnd causes that phase to turn off, allowing the ADP3194 to operate as a 2-, 3-, or 4-phase controller. 28 vcc a 300 resistor should be placed between the 12 v system supply and the vcc pin to ensure 5 v.
ADP3194 rev. 0 | page 7 of 32 typical performance characte ristic and test circuits frequency (khz) r t (k ? ) 06022-003 0 1000 2000 3000 4000 5000 6000 7000 0 100 200 300 400 500 600 700 800 900 figure 3. master cl ock frequency vs. r t (k) 250k? 12 v 1.25v 1f 100nf 100nf ADP3194 1 2 3 28 27 26 4 8 10 12 14 5 6 7 21 24 23 22 9 11 17 18 19 13 15 16 20 25 6-bit code 250k? 20k? 1k? 12nf + vcc pwm1 pwm2 pwm3 pwm4 sw1 sw2 sw3 sw4 gnd cscomp cssum csref ilimit vid4 vid3 vid2 vid1 vid0 vid5 fbrtn fb comp pwrgd en delay rt rampadj 06022-004 300? figure 4. closed-loop output voltage accuracy cssum 18 cscomp 17 28 vcc csref 16 gnd 19 39k ? 100nf 1k ? 1.0v ADP3194 v os = cscomp ? 1v 40 12 v 300 ? 06022-005 figure 5. current sense amplifier v os v fb = fb v = 80mv ? fb v =0mv cssum 18 cscomp 17 28 vcc csref 16 gnd 19 200k ? 100nf v 1.0v ADP3194 200k ? 12 v 300? 06022-006 figure 6. positioning voltage
ADP3194 rev. 0 | page 8 of 32 theory of operation the ADP3194 combines a multimode, fixed frequency pwm control with multiphase logic outputs for use in 2-, 3-, and 4-phase synchronous buck cpu core supply power converters. the internal vid dac is designed to interface with the intel 6-bit vrd/vrm 10- and 10.1-compatible cpus. multiphase operation is important for producing the high currents and low voltages demanded by todays microprocessors. handling the high currents in a single-phase converter places high thermal demands on the components in the system, such as the inductors and mosfets. the multimode control of the ADP3194 ensures a stable, high performance topology for ? balancing currents and thermals between phases ? high speed response at the lowest possible switching frequency and output decoupling ? minimizing thermal switching losses due to lower frequency operation ? tight load line regulation and accuracy ? high current output for up to 4-phase operation ? reduced output ripple due to multiphase cancellation ? pc board layout noise immunity ? ease of use and design due to independent component selection ? flexibility in operation for tailoring design to low cost or high performance startup sequence during startup, the number of operational phases and their phase relationship is determined by the internal circuitry that monitors the pwm outputs. normally, the ADP3194 operate as a 4-phase pwm controller. grounding the pwm4 pin programs 3-phase operation, and grounding the pwm3 pin and the pwm4 pin programs 2-phase operation. when the ADP3194 are enabled, the controller outputs a volt- age on pwm3 and pwm4, which is approximately 675 mv. an internal comparator checks each pins voltage vs. a threshold of 300 mv. if the pin is grounded, it is below the threshold, and the phase is disabled. the output resistance of the pwm pins is approximately 5 k during this detection time. any external pull-down resistance connected to the pwm pins should not be less than 25 k to ensure proper operation. pwm1 and pwm2 are disabled during the phase detection interval that occurs during the first two clock cycles of the internal oscillator. after this time, if the pwm output is not grounded, the 5 k resistance is removed and it switches between 0 v and 5 v. if the pwm output is grounded, it remains off. the pwm outputs are logic-level devices intended for driving external gate drivers, such as the adp3120a. because each phase is monitored inde- pendently, operation approaching 100% duty cycle is possible. also, more than one output can be on at the same time for overlapping phases. master clock frequency the clock frequency of the ADP3194 is set with an external resistor connected from the rt pin to ground. the frequency follows the graph in figure 3 . to determine the frequency per phase, the clock is divided by the number of phases in use. if pwm4 is grounded, divide the master clock by 3 for the fre- quency of the remaining phases. if pwm3 and pwm4 are grounded, divide by 2. if all phases are in use, divide by 4. output voltage differential sensing the ADP3194 differential sense compares a high accuracy vid dac and a precision reference to implement a low offset error amplifier. this maintains a worst-case specification of 9.5 mv differential sensing error over their full operating output voltage and temperature range. the output voltage is sensed between the fb pin and the fbrtn pin. connect fb through a resistor to the regulation point, usually the remote sense pin of the microprocessor. connect fbrtn directly to the remote sense ground point. the internal vid dac and precision reference are referenced to fbrtn, which has a minimal current of 100 a to allow accurate remote sensing. the internal error amplifier compares the output of the dac to the fb pin to regulate the output voltage. output current sensing the ADP3194 provide a dedicated current sense amplifier (csa) to monitor the total output current for proper voltage positioning vs. load current and for current-limit detection. sensing the load current at the output gives the total average current being delivered to the load, which is an inherently more accurate method than peak current detection or sampling the current across a sense element, such as the low-side mosfet. this amplifier can be configured several ways, depending on the objectives of the system: output inductor dcr sensing without a thermistor for lowest cost, output inductor dcr sensing with a thermistor for improved accuracy with tracking of inductor temperature, sense resistors for highest accuracy measurements.
ADP3194 rev. 0 | page 9 of 32 the positive input of the csa is connected to the csref pin, which is connected to the output voltage. the inputs to the amplifier are summed together through resistors from the sensing element (such as the switch node side of the output inductors) to the inverting input, cssum. the feedback resis- tor between cscomp and cssum sets the gain of the amplifier and a filter capacitor is placed in parallel with this resistor. the gain of the amplifier is programmable by adjusting the feedback resistor to set the load line required by the microprocessor. the current information is then given as the difference of csref ? cscomp. this difference signal is used internally to offset the vid dac for voltage positioning and as a differential input for the current-limit comparator. to provide the best accuracy for sensing current, the csa is designed to have a low offset input voltage. also, the sensing gain is determined by external resistors, so it can be made extremely accurate. active impedance control mode for controlling the dynamic output voltage droop as a function of output current, a signal proportional to the total output current at the cscomp pin can be scaled to equal the droop imped- ance of the regulator multiplied by the output current. this droop voltage is then used to set the input control voltage to the system. the droop voltage is subtracted from the dac reference input voltage directly to tell the error amplifier where the output voltage should be. this differs from previous imple- mentations and allows an enhanced feed-forward response. current-control mode and thermal balance the ADP3194 has individual inputs for each phase, which are used for monitoring the current in each phase. this informa- tion is combined with an internal ramp to create a current balancing feedback system, which has been optimized for initial current balance accuracy and dynamic thermal bal- ancing during operation. this current-balance information is independent of the average output current information used for positioning described previously. the magnitude of the internal ramp can be set to optimize the transient response of the system. it also monitors the supply voltage for feed-forward control for changes in the supply. a resistor connected from the power input voltage to the rampadj pin determines the slope of the internal pwm ramp. detailed information about programming the ramp is given in the application information section. external resistors can be placed in series with individual phases to create, if desired, an intentional current imbalance such as when one phase may have better cooling and can support higher currents. resistor r sw1 through resistor r sw4 (see the typical application circuit in figure 19 and figure 20) can be used for adjusting thermal balance. it is best to have the ability to add these resistors during the initial design, so make sure that place- holders are provided in the layout. to increase the current in any given phase, make r sw for this phase larger (make r sw = 0 for the hottest phase, and do not change during balancing). increasing r sw to only 500 makes a substantial increase in phase current. increase each r sw value by small amounts to achieve balance, starting with the coolest phase first. voltage control mode a high gain bandwidth voltage mode error amplifier is used for the voltage-mode control loop. the control input voltage to the positive input is set via the vid logic according to the voltages listed in table 4. this voltage is also offset by the droop voltage for active positioning of the output voltage as a function of the current, commonly known as active voltage positioning. the output of the amplifier is the comp pin, which sets the termi- nation voltage for the internal pwm ramps. the negative input (fb) is tied to the output sense location with a resistor (r b ) and is used for sensing and controlling the output voltage at this point. a current source from the fb pin flowing through r b is used for setting the no-load offset voltage from the vid voltage. the no-load voltage is negative with respect to the vid dac. the main loop compensation is incorporated into the feedback network between fb and comp. soft start the power-on ramp-up time of the output voltage is set with a capacitor and resistor in parallel from the delay pin to ground. the rc time constant also determines the current-limit latch-off time. in uvlo, or when en is logic low, the delay pin is held at ground. after the uvlo threshold is reached and en is logic high, the delay capacitor is charged with an internal 20 a current source. the output voltage follows the ramping voltage on the delay pin, limiting the inrush current. the soft start time depends on the value of the vid dac and c dly , with a secondary effect from r dly . see the application information section for detailed information on setting c dly . if en is taken low or if vcc drops below uvlo, the delay capacitor is reset to ground to be ready for another soft start cycle. figure 7 shows a typical soft start sequence for the ADP3194.
ADP3194 rev. 0 | page 10 of 32 06022-007 figure 7. typical start-up waveforms channel 1: pwrgd, channel 2: csref, channel 3: delay, channel 4: comp current-limit, short-circuit, and latch-off protection the ADP3194 compares a programmable current-limit setpoint to the voltage from the output of the current sense amplifier. the level of current limit is set with the resistor from the ilimit pin to ground. during normal operation, the voltage on ilimit is 3 v. the current through the external resistor is internally scaled to give a current-limit threshold of 10.4 mv/a. if the difference in voltage between csref and cscomp rises above the current- limit threshold, the internal current-limit amplifier controls the internal comp voltage to maintain the average output current at the limit. after the limit is reached, the 3 v pull-up on the delay pin is disconnected, and the external delay capacitor is discharged through the external resistor. a comparator monitors the delay voltage and shuts off the controller when the voltage drops below 1.8 v. the current-limit latch-off delay time is, therefore, set by the rc time constant discharging from 3 v to 1.8 v. the application information section discusses the selection of c dly and r dly . because the controller continues to cycle the phases during the latch-off delay time, the controller returns to normal operation if the short is removed before the 1.8 v threshold is reached. the recovery characteristic depends on the state of pwrgd. if the output voltage is within the pwrgd window, the controller resumes normal operation. however, if a short circuit has caused the output voltage to drop below the pwrgd threshold, a soft start cycle is initiated. the latch-off function can be reset by either removing and reap- plying vcc to the ADP3194 or by pulling the en pin low for a short time. to disable the short-circuit latch-off function, the external resistor to ground should be left open, and a high value (>1 m) resistor should be connected from delay to vcc. this prevents the delay capacitor from discharging, so the 1.8 v threshold is never reached. the resistor has an impact on the soft start time because the current through it adds to the internal 20 a current source. during startup, when the output voltage is below 200 mv, a secondary current limit is active. this is necessary because the voltage swing of cscomp cannot go below ground. this sec- ondary current limit controls the internal comp voltage to the pwm comparators to 2 v. this limits the voltage drop across the low-side mosfets through the current balance circuitry. an inherent per phase current limit protects individual phases, if one or more phases stops functioning because of a faulty com- ponent. this limit is based on the maximum normal mode comp voltage. 0 6022-008 figure 8. overcurrent latch-off waveforms channel 1: csref, channel 2: delay, channel 3: comp, channel 4: phase 1 switch node dynamic vid the ADP3194 has the ability to dynamically change the vid input while the controller is running. this allows the output voltage to change while the supply is running and supplying current to the load. this is commonly referred to as vid otf. a vid otf can occur under either light or heavy load conditions. the processor signals the controller by changing the vid inputs in multiple steps from the start code to the finish code. this change can be positive or negative. when a vid input changes state, the ADP3194 detects the change and ignores the dac inputs for a minimum of 400 ns. this time prevents a false code due to logic skew while the six vid inputs are changing. additionally, the first vid change initiates the pwrgd and crowbar blanking functions for a minimum of 100 s to prevent a false pwrgd or crowbar event. each vid change resets the internal timer.
ADP3194 rev. 0 | page 11 of 32 table 4. vid codes for the ADP3194 vid4 vid3 vid2 vid1 vid0 vid5 output 1 1 1 1 1 1 no cpu 1 1 1 1 1 0 no cpu 0 1 0 1 0 0 0.8375 v 0 1 0 0 1 1 0.8500 v 0 1 0 0 1 0 0.8625 v 0 1 0 0 0 1 0.8750 v 0 1 0 0 0 0 0.8875 v 0 0 1 1 1 1 0.9000 v 0 0 1 1 1 0 0.9125 v 0 0 1 1 0 1 0.9250 v 0 0 1 1 0 0 0.9375 v 0 0 1 0 1 1 0.9500 v 0 0 1 0 1 0 0.9625 v 0 0 1 0 0 1 0.9750 v 0 0 1 0 0 0 0.9875 v 0 0 0 1 1 1 1.0000 v 0 0 0 1 1 0 1.0125 v 0 0 0 1 0 1 1.0250 v 0 0 0 1 0 0 1.0375 v 0 0 0 0 1 1 1.0500 v 0 0 0 0 1 0 1.0625 v 0 0 0 0 0 1 1.0750 v 0 0 0 0 0 0 1.0875 v 1 1 1 1 0 1 1.1000 v 1 1 1 1 0 0 1.1125 v 1 1 1 0 1 1 1.1250 v 1 1 1 0 1 0 1.1375 v 1 1 1 0 0 1 1.1500 v 1 1 1 0 0 0 1.1625 v 1 1 0 1 1 1 1.1750 v 1 1 0 1 1 0 1.1875 v 1 1 0 1 0 1 1.2000 v vid4 vid3 vid2 vid1 vid0 vid5 output 1 1 0 1 0 0 1.2125 v 1 1 0 0 1 1 1.2250 v 1 1 0 0 1 0 1.2375 v 1 1 0 0 0 1 1.2500 v 1 1 0 0 0 0 1.2625 v 1 0 1 1 1 1 1.2750 v 1 0 1 1 1 0 1.2875 v 1 0 1 1 0 1 1.3000 v 1 0 1 1 0 0 1.3125 v 1 0 1 0 1 1 1.3250 v 1 0 1 0 1 0 1.3375 v 1 0 1 0 0 1 1.3500 v 1 0 1 0 0 0 1.3625 v 1 0 0 1 1 1 1.3750 v 1 0 0 1 1 0 1.3875 v 1 0 0 1 0 1 1.4000 v 1 0 0 1 0 0 1.4125 v 1 0 0 0 1 1 1.4250 v 1 0 0 0 1 0 1.4375 v 1 0 0 0 0 1 1.4500 v 1 0 0 0 0 0 1.4625 v 0 1 1 1 1 1 1.4750 v 0 1 1 1 1 0 1.4875 v 0 1 1 1 0 1 1.5000 v 0 1 1 1 0 0 1.5125 v 0 1 1 0 1 1 1.5250 v 0 1 1 0 1 0 1.5375 v 0 1 1 0 0 1 1.5500 v 0 1 1 0 0 0 1.5625 v 0 1 0 1 1 1 1.5750 v 0 1 0 1 1 0 1.5875 v 0 1 0 1 0 1 1.6000 v
ADP3194 rev. 0 | page 12 of 32 power good monitoring the power good comparator monitors the output voltage via the csref pin. the pwrgd pin is an open-drain output whose high level (when connected to a pull-up resistor) indicates that the output voltage is within the nominal limits specified in table 4 . these limits are based on the vid voltage setting. pwrgd goes low if the output voltage is outside of this specified range, if all of the vid dac inputs are high, or whenever the en pin is pulled low. pwrgd is blanked during a vid otf event for a period of 250 s to prevent false signals during the time the output is changing. the pwrgd circuitry also incorporates an initial turn-on delay time based on the delay ramp. the pwrgd pin is held low until the delay pin reaches 2.6 v. the time between when the pwrgd undervoltage threshold is reached and when the delay pin reaches 2.6 v provides the turn-on delay time. this time is incorporated into the soft start ramp. to ensure a 1 ms delay time on pwrgd, the soft start ramp must also be >1 ms. see the application information section for detailed information on setting c dly . output crowbar as part of the protection for the load and output components of the supply, the pwm outputs are driven low (turning on the low-side mosfets) when the output voltage exceeds the upper crowbar threshold. this crowbar action stops once the output voltage falls below the release threshold of approximately 550 mv. turning on the low-side mosfets pulls down the output as the reverse current builds up in the inductors. if the output over- voltage is due to a short in the high-side mosfet, this action current-limits the input supply or blows its fuse, protecting the microprocessor from being destroyed. output enable and uvlo for the ADP3194 to begin switching, the input supply (vcc) to the controller must be higher than the uvlo threshold, and the en pin must be higher than its logic threshold. if uvlo is less than the threshold or the en pin is logic low, the ADP3194 is disabled. this holds the pwm outputs at ground, shorts the delay capacitor to ground, and holds the ilimit pin at ground. in the application circuit, the ilimit pin should be connected to the od pins of the adp3120a drivers. grounding ilimit disables the drivers so that both the drvh and drvl are also grounded. this feature is important in preventing the discharge of the output capacitors when the controller is shut off. if the driver outputs were not disabled, a negative voltage could be generated during output due to the high current discharge of the output capacitors through the inductors.
ADP3194 rev. 0 | page 13 of 32 application information the design parameters for a typical intel vrd 10.1-compliant cpu application are as follows: ? input voltage (v in ) = 12 v ? vid setting voltage (v vid ) = 1.300 v ? duty cycle (d) = 0.108 ? nominal output voltage at no load (v onl ) = 1.281 v ? nominal output voltage at 101 a load (v ofl ) = 1.159 v ? static output voltage drop based on a 1.2 m load line (r o ) from no load to full load (v d ) = v onl ? v ofl = 1.281 v ? 1.159 v = 121.2 mv ? maximum output current (i o ) = 120 a ? maximum output current step (i o ) = 85 a ? number of phases (n) = 4 ? switching frequency per phase (f sw ) = 1.125 mhz setting the clock frequency the ADP3194 uses a fixed-frequency control architecture. the frequency is set by an external timing resistor (r t ). the clock frequency and the number of phases determine the switching frequency per phase, which relates directly to switching losses and the sizes of the inductors and/or the input and output capaci- tors. with n = 4 for four phases, a clock frequency of 4 mhz sets the switching frequency (f sw ) of each phase to 1 mhz, which represents a practical trade-off between the switching losses and the sizes of the output filter components. figure 3 shows that to achieve 4 mhz oscillator frequency, the correct value for r t is 84 k. 3 mhz oscillator frequency, the correct value for r t is 138 k. 2 mhz oscillator frequency, the correct value for r t is 247 k. alternatively, the value for r t can be calculated using ? = k79 pf6.4 3 sw t fn r (1) where 4.6 pf and 79 k are internal ic component values. for good initial accuracy and frequency stability, a 1% resistor is recommended. soft start and current-limit latch-off delay times because the soft start and current-limit latch-off delay functions share the delay pin, these two parameters must be considered together. the first step is to set c dly for the soft start ramp. this ramp is generated with a 20 a internal current source. the value of r dly has a second-order impact on the soft start time because it sinks part of the current source to ground. however, as long as r dly is kept greater than 200 k, this effect is minor. the value for c dly can be approximated by vid ss dly vid dly v t r v c ? ? ? ? ? ? ? ? ?= 2 a20 (2) where t ss is the desired soft start time. assuming an r dly of 390 k and a desired soft start time of 3 ms, c dly is 36 nf. the closest standard value for c dly is 39 nf. once c dly is chosen, r dly can be calculated for the current-limit latch- off time by dly delay dly c t r = 96.1 (3) if the result for r dly is less than 200 k, a smaller soft start time should be considered by recalculating equation 2, or a longer latch- off time should be used. r dly should never be less than 200 k. in this example, a delay time of 9 ms results in r dly = 452 k. the closest standard 5% value is 470 k. inductor selection the choice of inductance for the inductor determines the ripple current in the inductor. less inductance leads to more ripple current, which increases the output ripple voltage and conduction losses in the mosfets; but it allows using smaller inductors and, for a specified peak-to-peak transient deviation, less total output capacitance. conversely, a higher inductance means lower ripple current and reduced conduction losses but requires larger inductors and more output capacitance for the same peak-to-peak transient deviation. in any multiphase converter, a practical value for the peak-to-peak inductor ripple current is less than 50% of the maximum dc current in the same inductor. equation 4 shows the relationship between the inductance, oscillator frequency, and peak-to-peak ripple current in the inductor. ( ) lf d v i sw vid r ? = 1 (4) equation 5 can be used to determine the minimum inductance based on a given output ripple voltage. ( ) ( ) ripple sw o vid vf dnrv l ? 1 (5) solving equation 5 for a 10 mv p-p output ripple voltage yields () nh802 mv4.4hzm125.1 108.01 m 21.v1.3 = ? l if the resulting ripple voltage is less than it was designed for, make the inductor smaller until the ripple value is met. this allows optimal transient response and minimum output decoupling.
ADP3194 rev. 0 | page 14 of 32 the smallest possible inductor should be used to minimize the number of output capacitors. for this example, choosing a 280 nh inductor is a good starting point and gives a calculated ripple current of 3.68 a. the inductor should not saturate at the peak current of 31.84 a and should be able to handle the sum of the power dissipation caused by the average current of 30 a in the winding and core loss. designing an inductor once the inductance is known, the next step is either to design an inductor or to find a standard inductor that comes as close as possible to meeting the overall design goals. the first decision in designing the inductor is to choose the core material. several possibilities for providing low core loss at high frequencies include the powder cores (for example, kool-m? from magnetics, inc. or from micrometals) and the gapped soft ferrite cores (for example, 3f3 or 3f4 from philips). avoid low frequency powdered iron cores due to their high core loss, espe- cially when the inductor value is relatively low and the ripple current is high. the best choice for a core geometry is a closed-loop type such as a potentiometer core, pq, u, or e core or toroid. a good compromise between price and performance is a core with a toroidal shape. many useful magnetics design references are available for quickly designing a power inductor, such as magnetic designer software? by intusoft and designing magnetic components for high-frequency dc-dc converters , by william t. mclyman, kg magnetics, inc., isbn 1883107008. selecting a standard inductor power inductor manufacturers can provide design consulta- tion and deliver power inductors optimized for high power applications upon request. such manufacturers include coilcraft, coiltronics, sumida electric company, and vishay intertechnology. sense resistor a dedicated sense resistor can be used for current sensing. an advantage to this is the fact that there is much less temperature variation than using the dcr method. therefore, a thermistor is not required. the trade-off is that a sense resistor is required for each phase. so, one thermistor is saved, but four sense resistors are needed in a four phase design. also, there is extra power dissi- pation due to the sense resistor in series with the power delivery. sense resistor selection the resistance value of the sense resistor must be chosen to minimize the conduction loss, but be large enough for accurate current measurement. the lower the resistance, the lower the signal to noise ratio that appears at the ADP3194 input. this directly affects the current sense accuracy. a sense resistor of 1 m is chosen. the power loss in the resistor is calculated as: sense rs rip = 2 (6) if the design has 30 a per phase, then: mw900m1a30a30 = = rs p this results in a 900mw conduction loss through the sense resistor in a 30 a per phase design. therefore, a 1 m, 1 w sense resistor is chosen. there is a parasitic inductance (l p ) associated with the sense resistor. this value can be found on the data sheet of the sense resistor. a typical value is of the order of 2.2 nh. output droop resistanceCsense resistor the design requires the regulator output voltage measured at the cpu pins to drop when the output current increases. the specified voltage drop corresponds to a dc output resistance (r o ). the output current is measured by summing the voltage across each inductor and passing the signal through a low-pass filter. this summer filter is the cs amplifier configured with resistors r ph(x) (summers), and r cs and c cs (filter). the output resistance of the regulator is set by the following equations: () sense xph cs o r r r r = (7) cs sense p cs rr l c = (8) where r sense is the resistance of the sense resistor. the user has the flexibility of choosing either r cs or r ph(x) . it is best to select r cs equal to 100 k, and then solve for r ph(x) by rearranging equation 6. cs o sense xph r r r r = )( (9) k 5.82 k 100 m 2.1 m 0.1 )( == xph r next, use equation 8 to solve for c cs . pf220 k 100 m 0.1 nh2.2 = = cs c therefore, set r cs equal to 100 k, c cs equal to 220 pf, and r ph equal to 82.5 k.
ADP3194 rev. 0 | page 15 of 32 cssum 18 cscomp 17 28 vcc csref 16 ADP3194 c cb1 c cb2 keep this path as short as possible and well away from switch node lines. r cb1 r ph1 + + v cc (core) v cc (core) rtn l2 280nh sense 1m ? 1w 560f/4v n sanyo sepc series 06022-009 figure 9. using a sense resistor output offset the intel specification requires that at no load should the nomi- nal output voltage of the regulator be offset to a value lower than the nominal voltage corresponding to the vid code. the offset is set by a constant current source flowing out of the fb pin (i fb ) and flowing through r b . the value of r b can be found using equation 10: fb onl vid b i vv r (10) k22.1 a5.15 v281.1v3.1 b r the closest standard 1% resistor value is 1.21 k. design comparison trade-off between dcr and sense resistor cost the inductor dcr method requires a thermistor, which costs about $0.03. the r sense method requires an extra sense resistors for each phase. this costs about 4 $0.07 = $0.28. if it can also meet the intel accuracy specification of 25mv across the full load range, then it is the preferred method in vr applications. accuracy table 5 shows the accuracy results for a 4-phase vr10.1 applica- tion, using dcr method and sense resistor method. as can be seen, the sense resistor method improves the accuracy slightly. however, since the dcr method meets the intel specification, it is the preferred solution, for cost reasons. table 5. input parameters design inputs sense resistor method dcr method parameter value value n 4 4 dcr 1.00 m 1.00 m loadline 1.00 m 1.00 m imax 120.0 a 120.0 a istep 90.0 a 90.0 a l error 0% 20% c error 5% 5% r sense error 1% 5% temp rise 70c 70c r sense vs. temp 0.0 % 26.6% no-load offset 19.0 mv 19.0 mv total output vripple 8.0 mv 8.0 mv inductor iripple 6.0 a 6.0 a gain factor for tc 12 12 30 28 26 24 22 20 18 16 14 0 20406080100120 output current (a) error (mv) error from loadline ( amount shown) this chart is active. vrd10.1 spec sense resistor method dcr method 06022-010 figure 10. accuracy comparison of dcr and sense resistor methods
ADP3194 rev. 0 | page 16 of 32 c out selection the required output decoupling for the regulator is typically recommended by intel for various processors and platforms. also, to determine what is required, use some simple design guidelines that are based on having both bulk and ceramic capacitors in the system. the first thing is to select the total amount of ceramic capaci- tance. this is based on the number and type of capacitor to be used. the best location for ceramic capacitors is inside the socket, with 12 to 18 of size 1206 being the physical limit. additional ceramic capacitors can be placed along the outer edge of the socket as well. combined ceramic values of 200 f to 400 f are recommended, usually made up of multiple 10 f or 22 f capacitors. select the number of ceramic capacitors, and find the total ceramic capacitance (c z ). next, there is an upper limit imposed on the total amount of bulk capacitance (c x ) when considering the vid otf voltage stepping of the output (voltage step v v in time t v with error of v err ). a lower limit is based on meeting the capacitance for load release for a given maximum load step, ? i o , and a max- imum allowable overshoot. the total amount of load release voltage is given as ' v o = ' i o u r o + ' v rl where ' v rl is the maximum allowable overshoot voltage. u ' ' u u t z vid o rl o o minx c v i v rn il c (11) z o v vid v vid v 2 o 2 maxx c l nkr v v t v v rnk l c uuu 1 1 2 )( (12) v err v v nk 1where to meet the conditions of these equations and transient response, the esr of the bulk capacitor bank (r x ) should be less than two times the droop resistance (r o ). if the c x(min) is larger than c x(max) , the system cannot meet the vid otf specification and may require the use of a smaller inductor or more phases (and may need the switching frequency to increase to keep the out-put ripple the same). this example uses 18, 22 f 1206 mlc capacitors ( c z = 396 f). the vid on-the-fly step change is 450 mv in 230 s with a settling error of 2.5 mv. the maximum allowable load release overshoot for this example is 50 mv, so solving for the bulk capacitance yields mf16.2f396 v3.1 85 mv50 m 2.14 a85nh280 )( u u u d a c minx mf5.40f3961 nh280mv450 m 2.16.44v3.1s250 1 v3.1 m 2.16.44 mv450nh280 2 2 2 )( u uuuu u u uu u d maxx c where k = 4.6. using four 560 f al-poly capacitors with a typical esr of 5 m each yields c x = 2.24 mf with an r x = 1.25 m. one last check should be made to ensure that the esl of the bulk capacitors (l x ) is low enough to limit the high frequency ringing during a load change. this is tested using nh14.12 m 2.1f396 2 2 2 u ud uud x o z x l qrcl (13) where q is limited to the square root of 2 to ensure a critically damped system. in this example, l x is approximately 175 ph for the four a1-polys capacitors, which satisfies this limitation. if the l x of the chosen bulk capacitor bank is too large, the number of ceramic capaci- tors may need to be increased if there is excessive ringing. for this multimode control tech nique, all ceramic designs can be used as long as the conditions of equation 11, equation 12, and equation 13 are satisfied.
ADP3194 rev. 0 | page 17 of 32 ramp resistor selection the ramp resistor (r r ) is used for setting the size of the internal pwm ramp. the value of this resistor is chosen to provide the best combination of thermal balance, stability, and transient response. the following equation is used for determining the optimum value: k 118 pf 5 m 33 . 6 5 3 nh 0 28 0.2 3 = = = r r ds d r r r c r a l a r (14) where: a r is the internal ramp amplifier gain a d is the current balancing amplifier gain r ds is the total low-side mosfet on resistance c r is the internal ramp capacitor value. the internal ramp voltage magnitude can be calculated by using () () v v f c r v d a v r sw r r vid r r m 0 5 3 mhz 125 . 1 pf 5 k 118 v 1.3 0.108 1 0.2 1 = ? = ? = (15) the size of the internal ramp can be made larger or smaller. if it is made larger, stability and transient response improve, but thermal balance degrades. likewise, if the ramp is made smaller, thermal balance improves at the sacrifice of transient response and stability. the factor of 3 in the denominator of equation 14 sets a ramp size that gives an optimal balance for good stability, transient response, and thermal balance. comp pin ramp a ramp signal on the comp pin is due to the droop voltage and output voltage ramps. this ramp amplitude adds to the internal ramp to produce the following overall ramp signal at the pwm input: () ? ? ? ? ? ? ? ? ? ? = o x sw r rt r c f n d n v v 1 2 1 (16) in this example, the overall ramp signal is 390 mv. current-limit setpoint to select the current-limit setpoint, first find the resistor value for r lim . the current-limit threshold for the ADP3194 is set with a 3 v source (v lim ) across r lim with a gain of 10.4 mv/a (a lim ). r lim can be found using o lim lim lim lim r i v a r = (17) for values of r lim greater than 500 k, the current limit can be lower than expected, so some adjustment of r lim may be needed. here, i lim is the average current limit for the output of the supply. in this example, choosing a peak current limit of 185 a for i lim results in r lim = 140 k. the limit of the per-phase current-limit described earlier is determined by () () 2 r max ds d bias r max comp phlim i r a v v v i + ? ? ? (18) for the ADP3194, the maximum comp voltage (v comp(max) ) is 3.3 v, the comp pin bias voltage (v bias ) is 1.2 v, and the current- balancing amplifier gain (a d ) is 5. using v r of 0.35 v and r ds(max) of 7 m, the per-phase peak current limit is calculated to be 51.8 a. although this number may seem high, this current level can be reached only with an absolute short at the output, and the current-limit latch-off function shuts down the regulator before overheating can occur. this limit can be adjusted by changing the ramp voltage (v r ), but make sure not to set the per-phase limit lower than the average per-phase current (i lim /n). the per-phase initial duty cycle limit is determined by () rt bias max comp max v v v d d ? = (19) in this example, the maximum duty cycle is 0.46. feedback loop comp ensation design optimized compensation of the ADP3194 allows the best pos- sible response of the regulators output to a load change. the basis for determining the optimum compensation is to make the regulator and output decoupling appear as an output imped- ance that is entirely resistive over the widest possible frequency range, including dc, and equal to the droop resistance (r o ). with the resistive output impedance, the output voltage droops in proportion to the load current at any load current slew rate. this ensures optimal positioning and allows minimization of the output decoupling.
ADP3194 rev. 0 | page 18 of 32 with the multimode feedback structure of the ADP3194, the feedback compensation must be set to make the converters output impedance, working in parallel with the output decoup- ling, to meet this goal. several poles and zeros created by the output inductor and decoupling capacitors (output filter) need to be compensated for. a type-three compensator on the voltage feedback is adequate for proper compensation of the output filter. equation 20 to equation 28 yield an optimal starting point for the design; some adjustments may be necessary to account for pcb and component parasitic effects (see the layout and component placement section). the first step is to compute the time constants for all of the poles and zeros in the system: () vid o x rt vid rt sense ds d o e v r c n v d n l v v r r a r n r ? + + + = 1 2 (20) () x o o x o x a r r r r l r r c t ? + ? = (21) ( ) x o x b c r r r t ? + = (22) e vid sw ds d rt c r v f r a l v t ? ? ? ? ? ? ? ? ? = 2 (23) () o z o x o z x d r c r r c r c c t + ? = ' 2 (24) where: r' is the pcb resistance from the bulk capacitors to the ceramics r ds is the total low-side mosfet on resistance per phase. in this example, a d is 5, v rt equals 0.39 v, r' is approximately 0.5 m (assuming a 4-layer, 1 ounce motherboard), and l x is 175 ph for the four al-poly capacitors. the compensation values can then be solved using the following equations: b e a o a r r t r n c = (25) a c a c t r = (26) b b b r t c = (27) a d fb r t c = (28) these are the starting values, prior to tuning the design, to account for layout and other parasitic effects (see the layout and component placement section). the final values selected after tuning are c a = 3.3 nf, r a = 7.32 k, c b = 1 nf, c fb = 33 pf.
ADP3194 rev. 0 | page 19 of 32 figure 11 and figure 12 show the typical transient response using these compensation values. 06022-011 figure 11. typical transient response for design example load step 06022-012 figure 12. typical transient response for design example load release c in selection and input current di/dt reduction in continuous inductor current mode, the source current of the high-side mosfet is approximately a square wave with a duty ratio equal to n v out /v in and an amplitude of one- n th the maximum output current. to prevent large voltage transients, a low esr input capacitor, sized for the maximum rms current, must be used. the maximum rms capacitor current is given by a14.71 0.1084 1 a191108.0 1 1 =? = ? = crms o crms i dn idi (29) the capacitor manufacturers ripple current ratings are often based on only 2000 hours of life. this makes it advisable to further derate the capacitor or to choose a capacitor rated at a higher temperature than required. several capacitors can be placed in parallel to meet size or height requirements in the design. in this example, the input capacitor bank is formed by two 2700 f, 16 v aluminum electrolytic capacitors and eight 4.7 f ceramic capacitors. to reduce the input current di/dt to a level below the recom- mended maximum of 0.1 a/s, an additional small inductor (l > 370 nh at 18 a) should be inserted between the converter and the supply bus. this inductor also acts as a filter between the converter and the primary power source. 06022-012 figure 13. efficiency of the circuit of figure 10 vs. output current
ADP3194 rev. 0 | page 20 of 32 tuning the ADP3194 1. build a circuit based on the compensation values computed from the equations used in the example. 2. hook up the dc load to circuit, turn it on, and verify its operation. also, check for jitter at no load and full load. dc load line setting 3. measure the output voltage at no load (v nl ). verify it is within tolerance. 4. measure the output voltage at full load cold (v flcold ). let the board sit for ~10 minutes at full load, and then measure the output (v flhot ). if there is a change of more than a few millivolts, adjust r cs1 and r cs2 , using equation 30 and equation 31. () () flhot nl flcold nl oldcs2 newcs2 vv v v r r ? ? = (30) 5. repeat step 4 until the cold and hot voltage measurements remain the same. 6. measure the output voltage from no load to full load, using 5 amps steps. compute the load line slope for each change, and then average to get the overall load line slope (r omeas ). 7. if r omeas is off from r o by more than 0.05 m, use the following to adjust the r ph values: () () o omeas oldph newph r r r r = (31) 8. repeat step 6 and step 7 to check the load line, and repeat adjustments if necessary. 9. once the dc load line adjustment is complete, do not change r ph , r cs1 , r cs2 , or r th for the remainder of the procedure. 10. measure the output ripple at no load and full load with a scope, and make sure it is within specifications. () () () () () () () () () () () () c25 c25 c25 c25 1 1 ? ? ? + + = th th oldcs1 newcs2 oldcs1 th oldcs1 th oldcs1 newcs1 r rr rr r r r r r (32)
ADP3194 rev. 0 | page 21 of 32 ac load line setting 11. remove the dc load from the circuit and hook up the dynamic load. 12. hook up the scope to the output voltage and set it to dc coupling, with the time scale at 100 s/div. 13. set the dynamic load for a transient step of about 40 a at 1 khz with a 50% duty cycle. 14. measure the output waveform (if not visible, use dc offset on scope to view). try to use a vertical scale of 100 mv/div or finer. this waveform should look similar to figure 14 . v acdrp v dcdrp 06022-014 figure 14. ac load line waveform 15. use the horizontal cursors to measure v acdrp and v dcdrp , as shown in figure 14. do not measure the undershoot or overshoot that happens immediately after this step. if v acdrp and v dcdrp are different by more than a few milli- volts, use equation 38 to adjust c cs . it may be necessary to parallel different values to get the correct one, because there are limited standard capacitor values available. it is a good idea to have locations for two capacitors in the layout for this. () () dcdrp acdrp oldcs newcs v v c c = (38) 16. repeat step 11 to step 13, and repeat the adjustments, if necessary. once complete, do not change c cs for the remainder of the procedure. 17. set the dynamic load step to maximum step size. do not use a step size larger than needed, and verify that the output waveform is square, which means that v acdrp and v dcdrp are equal. initial transient setting 18. with the dynamic load still set at the maximum step size, expand the scope time scale to see 2 s/div to 5 s/div. the waveform may have two overshoots and one minor under- shoot (see figure 15 ). here, v droop is the final desired value. v droop v tran1 v tran2 06022-015 figure 15. transient setting waveform 19. if both overshoots are larger than desired, try making the following adjustments: make the ramp resistor larger by 25% (r ramp ). for v tran1 , increase c b , or increase the switching frequency. for v tran2 , increase r a , and decrease c a by 25%. if these adjustments do not change the response, the output decoupling is the limiting factor. check the output response every time a change is made, or nodes are switched, to make sure the response remains stable. 20. for load release (see figure 16 ), if v tranrel is larger than v tran1 (see figure 15 ), there is not enough output capaci- tance. either more capacitance is needed or the inductor values need to be smaller. if inductors are changed, start the design again using the spreadsheet and this tuning procedure. v droop v tranrel 06022-016 figure 16. transient setting waveform
ADP3194 rev. 0 | page 22 of 32 because the ADP3194 turns off all of the phases (switches induc- tors to ground), there is no ripple voltage present during load release. thus, headroom does not need to be added for ripple, allowing load release (v tranrel ) to be larger than v tran1 by the amount of ripple and still meet specifications. if v tran1 and v tranrel are less than the desired final droop, this implies that capacitors can be removed. when removing capaci- tors, also check the output ripple voltage to make sure it is still within specifications. rampadj filter it is recommended that a filter be placed on the rampadj line. on the ADP3194, the vcc is 5 v, but the rampadj still needs to be connected to the 12 v input supply. therefore, the filter is needed to remove noise from the 12 v input supply. a 1 k resistor and 1 f cap are recommended for this filter. shunt resistor design when replacing an existing adp3181 design with the ADP3194, the shunt resistor value needs to be determined. a trade-off can be made between the power dissipated in the shunt resistor and the uvlo threshold. figure 17 shows the typical resistor value needed to realize certain uvlo voltages. it also gives the maxi- mum power dissipated in the shunt resistor for these uvlo voltages. the maximum power dissipated is calculated using equation 33. () () ( ) shunt min cc max in max r v v p 2 ? = (33) where: v in(max) is the maximum voltage from the 12 v input supply. (if the 12 v input supply is 12 v 5%, then v in(max) = 12.6 v. if the 12 v input supply is 12 v 10%, then v in(max) = 13.2 v.) figure 17 shows the power when v in(max) = 12.6 v. v cc(min) is the minimum v cc voltage of the ADP3194. it is specified as 4.75 v. r shunt is the shunt resistor value. the cecc standard specification for power rating in surface mount resistors is 0603 = 0.1 w, 0805 = 0.125 w, 1206 = 0.25 w. for example, uvlo voltage specification = 8 v. from figure 17, a shunt resistor value of 420 is recommended. from figure 17, the power dissipation is 140 mw. the user can choose any of the following: two 840 , 0603 resistors in parallel two 840 , 0805 resistors in parallel one 420 , 1206 resistor. r shunt ( ? ) uvlo (v) power (mw) 100 6.5 9.5 9.0 8.5 8.0 7.5 7.0 0.10 0.40 0.35 0.30 0.25 r shunt p shunt 0.20 0.15 700 600 500 400 300 200 06022-017 figure 17. typical shunt resistor value and power dissipation for different uvlo voltages
ADP3194 rev. 0 | page 23 of 32 design example using dcr method the following are guidelines for designs that use an inductor dcr method instead of sense resistor method. inductor selection using dcr use the method and equations described in the inductor selection section to calculate the inductor. an important factor in the inductor design is the dcr, which is used for measuring the phase currents. a large dcr can cause excessive power losses, while too small a value can lead to increased measurement error. a good rule is to have the dcr be about 1 to 1? times the droop resistance (r o ). for this design, an inductor with a dcr of 1.4 m is used. designing an inductor using dcr once the inductance and dcr are known, the next step is either to design an inductor or to find a standard inductor that comes as close as possible to meeting the overall design goals. it is also important to have the inductance and dcr tolerance specified to control the accuracy of the system. 15% inductance and 8% dcr (at room temperature) are reasonable tolerances most manufacturers can meet. inductor dcr temperature correction with the inductors dcr being used as the sense element and copper wire being the source of the dcr, compensation is needed for temperature changes of the inductors winding. fortunately, copper has a well-known temperature coefficient (tc) of 0.39%/c. if r cs is designed to have an opposite and equal percentage change in resistance to that of the wire, it cancels the tempera- ture variation of the inductors dcr. due to the nonlinear nature of ntc thermistors, resistor r cs1 and resistor r cs2 are needed. see figure 18 to linearize the ntc and produce the desired temperature tracking. cssum 18 cscomp place as close as possible to nearest inductor or low-side mosfet 17 csref 16 ADP3194 c cs1 c cs2 r cs1 r th r cs2 keep this path as short as possible and well away from switch node lines to switch nodes to v out sense r ph1 r ph3 r ph2 0 6022-018 figure 18. temperature compensation circuit values the following procedure and equations yield values to use for r cs1 , r cs2 , and r th (the thermistor value at 25c) for a given r cs value. select an ntc based on type and value. because there is not a value yet, start with a thermistor with a value close to r cs . the ntc should also have an initial tolerance of better than 5%. based on the type of ntc, find its relative resistance value at two temperatures. the temperatures that work well are 50c and 90c. these resistance values are called a (r th(50c) /r th(25c) ) and b (r th(90c) /r th(25c) ). the ntcs relative value is always 1 at 25c. find the relative values of r cs required for each of these temperatures. this is based on the percentage change needed, which in this example is initially 0.39%/c. these are called r 1 (1/(1 + tc ( t 1 ? 25))) and r 2 (1/(1 + tc ( t 2 ? 25))), where tc = 0.0039 for copper. t 1 = 50c and t 2 = 90c are chosen. from this, calculate that r 1 = 0.9112 and r 2 = 0.7978. compute the relative values for r cs2 , r cs1 , and r th using ( ) () ( ) () () ( ) barabrba rabrbarrba r 2 1 1 2 21 cs2 ????? ? +? ? ? = 1 1 1 1 (34) ( ) cs2 1 cs2 cs1 rr a r a r ? ? ? ? = 1 1 1 (35) cs1 cs2 th rr r 1 1 1 1 ? ? = (36) calculate r th = r th r cs , then select the closest value of thermis- tor available. also, compute a scaling factor k based on the ratio of the actual thermistor value used relative to the computed one: () () calculated th actual th r r k = (37) calculate values for r cs1 and r cs2 using equation 38 and equation 39: cs1 cs cs1 rkrr = (38) ( ) ( ) ( ) cs2 cs cs2 rkkrr + ? = 1 (39) for this example, r cs has been calculated to be 110 k. start with a thermistor value of 100 k. next, look through the available 0603-size thermistors, and find a vishay nths0603n01n1003jr ntc thermistor with a = 0.3602 and b = 0.09174. from these, compute r cs1 = 0.3795, r cs2 = 0.7195, and r th = 1.075. solve for r th , which yields 118.28 k. then, choose 100 k, which makes k = 0.8455. finally, r cs1 and r cs2 are 35.3 k and 83.9 k. choose the closest 1% resistor values, which yield a choice of 35.7 k or 84.5 k.
ADP3194 rev. 0 | page 24 of 32 output droop resistanceCdcr method the design requires the regulator output voltage measured at the cpu pins to drop when the output current increases. the specified voltage drop corresponds to a dc output resistance (r o ). the output current is measured by summing the voltage across each inductor and passing the signal through a low-pass filter. this summer filter is the cs amplifier configured with r ph(x) (summers), r cs , and c cs (filter). the output resistance of the regulator is set by the following equations: () l xph cs o r r r r = (40) cs l cs rr l c = (41) where r l is the dcr of the output inductors. the user has the flexibility of choosing either r cs or r ph(x) . it is best to select r cs (equal to 100 k) and then solve for r ph(x) by rearranging equation 6. () cs o l x ph r r r r = (42) () k 5.82 k 100 m 2.1 m 0.1 == xph r next, use equation 41 to solve for c cs . nf8.2 k100m0.1 nh280 = = cs c it is best to have a dual location for c cs in the layout, so that standard values can be used in parallel to get as close as possible to the value desired. for accuracy, c cs should be a 5% or 10% npo capacitor. this example uses a 5% combination for c cs of 2.2 nf and 560 pf in parallel. power mosfets this section is only applicable if power mosfets need to be selected. for this example, the n-channel power mosfets have been selected for one high-side switch and two low-side switches per phase. the main selection parameters for the power mosfets are v gs(th) , q g , c iss , c rss , and r ds(on) . the minimum gate drive voltage (the supply voltage to the adp3120a) dictates whether standard threshold or logic-level threshold mosfets must be used. with v gate ~10 v, logic-level threshold mosfets (v gs(th) < 2.5 v) are recommended. the maximum output current (i o ) determine1s the r ds(on) requirement for the low-side (synchronous) mosfets. with the ADP3194, currents are balanced between phases, thus the current in each low-side mosfet is the output current divided by the total number of mosfets (n sf ). with conduction losses being dominant, the following equation shows the total power being dissipated in each synchronous mosfet in terms of the ripple current per phase (i r ) and average total output current (i o ): () () sfds sf r sf o sf r n in n i dp ? ? ? ? ? ? ? ? ? ? ? ? ? ? ? ? + ? ? ? ? ? ? ? ? ?= 2 2 12 1 1 (43) knowing the maximum output current being designed for and the maximum allowed power dissipation, it is possible to find the required r ds(on) for the mosfet. for d-pak mosfets up to an ambient temperature of 50c, a safe limit for p sf is 1 w to 1.5 w at 120c junction temperature. thus, for this example (119 a maximum), r ds(sf) (per mosfet) < 7.5 m. this r ds(sf) is also at a junction temperature of about 120c, so be certain to account for this temperature when making this selection. this example uses two lower-side mosfets at 4.8 m each at 120 c. another important factor for the synchronous mosfet is the input capacitance and feedback capacitance. the ratio of the feedback to input needs to be small (less than 10% is recom- mended) to prevent accidental turn-on of the synchronous mosfets when the switch node goes high. also, the time to switch the synchronous mosfets off should not exceed the nonoverlap dead time of the mosfet driver (40 ns typical for the adp3120a). the output impedance of the driver is approximately 2 , ? and the typical mosfet input gate resistances are about 1 ? to 2 , so a total gate capacitance of less than 6000 pf should be adhered to. because there are two mosfets in parallel, the input capacitance for each synchronous mosfet should be limited to 3000 pf. the high-side (main) mosfet has to be able to handle two main power dissipation components: conduction and switching losses. the switching loss is related to the amount of time it takes for the main mosfet to turn on and off and to the current and the voltage that are being switched. basing the switching speed on the rise and fall time of the gate driver impedance and mosfet input capacitance, the follow- ing equation provides an approximate value for the switching loss per main mosfet: () iss mf g m f occ sw mf s c n n r n i v f p = 2 (44) where: n mf is the total number of main mosfets, r g is the total gate resistance (2 for the adp3120a and about 1 for typical high speed switching mosfets, making r g = 3 ), c iss is the input capacitance of the main mosfet. adding more main mosfets ( n mf ) does not really help the switching loss per mosfet because the additional gate capaci- tance slows switching. the best way to reduce switching loss is to use lower gate capacitance devices.
ADP3194 rev. 0 | page 25 of 32 the conduction loss of the main mosfet is given by the following equation: () () mf ds mf r mf mf c r n i n n d p ? ? ? ? ? ? ? ? ? ? ? ? ? ? ? ? + ? ? ? ? ? ? ? ? = 2 2 o 12 1 i (45) where r ds(mf) is the on resistance of the mosfet. typically, for main mosfets, the highest speed (low c iss ) device is preferred, but these usually have higher on resistance. select a device that meets the total power dissipation (about 1.5 w for a single d-pak) when combining the switching and conduction losses. for this example, an ntd40n03l was selected as the main mosfet (eight total; n mf = 8), with a c iss = 584 pf (maximum) and r ds(mf) = 19 m (maximum at t j = 120c). an ntd110n02l was selected as the synchronous mosfet (eight total; n sf = 8), with c iss = 2710 pf (maximum) and r ds(sf) = 4.8 m (maximum at t j = 120c). the synchronous mosfet c iss is less than 3000 pf, satisfying that requirement. solving for the power dissipation per mosfet at io = 119 a and ir = 11 a yields 958 mw for each synchronous mosfet and 872 mw for each main mosfet. these numbers comply with the guideline to limit the power dissipation to 1 w per mosfet. one last thing to consider is the power dissipation in the driver for each phase. this is best described in terms of the q g for the mosfets and is given by the following equation: () cc cc gsf sf gmf mf sw drv v i q n q n n f p ? ? ? ? ? ? ? ? + + = 2 (45) where: q gmf is the total gate charge for each main mosfet q gsf is the total gate charge for each synchronous mosfet the standby dissipation factor for the driver is i cc v cc . for the adp3120a, the maximum dissipation should be less than 400 mw. in this example (with i cc = 7 ma, q gmf = 5.8 nc, and q gsf = 48 nc) 297 mw is found in each driver, which is below the 400 mw dissipation limit. see the adp3120a data sheet for more details.
ADP3194 rev. 0 | page 26 of 32 layout and component placement the following guidelines are recommended for optimal per- formance of a switching regulator in a pc system. general recommendations for good results, a pcb with at least four layers is recom- mended. this allows the needed versatility for control circuitry interconnections with optimal placement; power planes for ground, input, and output power; and wide inter- connection traces in the remainder of the power delivery current paths. each square unit of 1 ounce copper trace has a resistance of ~0.53 m at room temperature. whenever high currents must be routed between pcb layers, vias should be used liberally to create several parallel current paths. then, the resistance and inductance introduced by these current paths is minimized, and the via current rating is not exceeded. if critical signal lines, including the output voltage sense lines of the ADP3194, must cross through power circuitry, it is best if a signal ground plane can be interposed between those signal lines and the traces of the power circuitry. this serves as a shield to minimize noise injection into the signals at the expense of making the signal ground noisier. use an analog ground plane around and under the ADP3194 as a reference for the components associated with the controller. this plane should be tied to the nearest output decoupling capacitor ground and not tied to any other power circuitry. this prevents power currents from flowing in the ground plane. locate the components around the ADP3194 close to the con- troller with short traces. the most important traces to keep short, and away from other traces, are the fb pin and the cssum pin. connect the output capacitors as close as possible to the load (or connector), for example, a microprocessor core that receives the power. if the load is distributed, the capacitors should also be distributed and generally be in proportion to where the load tends to be more dynamic. avoid crossing any signal lines over the switching power path loop, as described in the power circuitry recommendations section. power circuitry recommendations the switching power path should be routed on the pcb to encom- pass the shortest possible length in order to minimize radiated switching noise energy (that is, emi) and conduction losses in the board. failure to take proper precautions often results in emi problems for the entire pc system as well as noise-related operational problems in the power converter control circuitry. the switching power path is the loop formed by the current path through the input capacitors and the power mosfets, including all interconnecting pcb traces and planes. using short and wide interconnection traces is especially critical in this path for two reasons: it minimizes the inductance in the switching loop, which can cause high energy ringing; and it accommodates the high current demand with minimal voltage loss. whenever a power dissipating component, (for example, a power mosfet), is soldered to a pcb, the liberal use of vias, both directly on the mounting pad and immediately surrounding it, is recommended. this improves current rating through the vias and also improves thermal performance from vias extended to the opposite side of the pcb, where a plane can more readily transfer the heat to the air. make a mirror image of any pad being used to heat-sink the mosfets on the opposite side of the pcb to achieve the best thermal dissipation to the air around the board. to further improve thermal performance, use the largest possible pad area. the output power path should also be routed to encompass a short distance. the output power path is formed by the current path through the inductor, the output capacitors, and the load. for best emi containment, a solid power ground plane should be used as one of the inner layers extending fully under all the power components. signal circuitry recommendations the output voltage is sensed and regulated between the fb pin and the fbrtn pin, which connect to the signal ground at the load. to avoid differential-mode noise pickup in the sensed signal, the loop area should be small. thus, the fb and fbrtn traces should be routed adjacent to each other on top of the power ground plane back to the controller. the feedback traces from the switch nodes should be connected as close as possible to the inductor. the csref signal should be kelvin connected through a 10 resistor to the center point of the copper bar, which is the v core common node for the inductors of all the phases (see figure 19 and figure 20 ).
ADP3194 rev. 0 | page 27 of 32 12vr od# pwm1 pwm2 pwm4 pwm3 vcc1 cs_ph1 cs_ph2 cs_ph4 cs_ph3 12vr vsssocket vccsense vsssense vcc1 vccsocket vid2 vid3 pwm1 pwm4 vid4 vid5 vid0 pwm3 vid1 pwm2 phase1 phase2 phase3 phase4 +1 2v csref gnd gnd vccp d6 bzx84c5v1fsct tp23 csref 1 + ce13 100uf/16v 5x11mm 1 2 r ph2 15 k 1% 1 2 tp10 pwm2 1 r13 715 ohm 1 2 c b 1.5 nf 1 2 tp 7 od# 1 r th 100kohm thermistor 5% 1 2 12 r33 220k 1 2 tp18 agnd 1 r21 0 1 2 c a 1 nf x7 r 1 2 tp 3 comp 1 jp3 1 2 3 4 r2 0 ni 1 2 r16 0 1 2 c23 1uf/16v x7r(0805 ) 1 2 jp1 shortpin 1 2 sw1 r ph3 15 k 1% 1 2 c44 100 0p 1 2 tp13 vcc 1 tp 4 pwrgd 1 c dly 39n 1 2 tp11 pwm3 1 r35 1k 1 2 r lim 140k 1% 1 2 d1 mm sd414 8 sod-123 1 2 r shunt 221 1 2 q23 nfet 1 3 2 rn1g 1k 7 8 r10 10 1 2 c24 10 0p npo 1 2 r11 10 1 2 r ph4 15 k 1% 1 2 c fb 15 pf 1 2 r cs1 01% 1 2 r44 r rn2g 1.5k 7 8 c43 39n x7 r 1 2 tp 2 vccp 1 dl1 1 grn tp6 delay 1 r r 118k 1% 1 2 tp12 pwm4 1 r45 r rn3g 3.9k 7 8 r34 0 1 2 r sc2 18.2k 1% 1 2 r1 0 1 2 tp 8 cscomp 1 r46 r tp20 fb 1 tp19 gnd 1 tp 9 pwm1 1 r b 1.21k 1% 1 2 r30 10 1 2 r14 105 ohm 1 2 r47 r r3 0 ni 1 2 r4 0 1 2 c sc2 npo 5% dni 1 2 r dly 470k 1 2 c46 10nf 1 2 c45 1uf 1 2 r ph1 15 k 1% 1 2 r a 7.32k 1% 1 2 r31 10 1 2 r32 470k 1 2 u1 ADP3194 1 2 3 4 5 6 7 8 9 10 11 12 13 14 28 27 26 25 24 23 22 21 20 19 18 17 16 15 vid4 vid3 vid2 vid1 vid0 vid5 fbrtn fb comp pwrgd en delay rt rampadj vcc pwm1 pwm2 pwm3 pwm4 sw1 sw2 sw3 sw4 gnd cscomp cssum csref ilimit r15 619 ohm 1 2 r t 154k 1% 1 2 c sc1 150 pf npo 5% 1 2 tp 5 en 1 tp 1 vssp 1 vsssocket vsssense vccsense vccsocket vtt_pwrg d vid0 vid1 vid2 vid4 cpu_vid5 phase_1_rdy phase_2_rdy phase_3_rdy phase_4_rdy vid3 cpu_vid3 cpu_vid4 cpu_vid2 vid5 cpu_vid1 cpu_vid0 +1 2v +1 2v rn1a 1k 1 14 j2 header_6x2 2 4 6 8 10 12 1 3 5 7 9 11 rn2d 1.5k 4 11 r41 374 ohm dl9 grn rn3e 3.9 k 5 10 q3 nfet 1 3 2 rn1b 1k 2 13 rn2a 1.5k 1 14 dl2 red dl1 0 grn rn3f 3.9 k 6 9 r42 374 ohm q4 nfet 1 3 2 rn2b 1.5k 2 13 q1 nfet 1 3 2 rn1c 1k 3 12 rn2c 1.5k 3 12 r43 374 ohm dl3 red q5 nfet 1 3 2 rn3d 3.9 k 4 11 rn2e 1.5k 5 10 rn1d 1k 4 11 rn1e 1k 5 10 rn2f 1.5k 6 9 dl4 red q6 nfet 1 3 2 rn1f 1k 6 9 dl1 red rn3a 3.9 k 1 14 dl5 red dl7 grn rn3b 3.9 k 2 13 r40 374 ohm q2 nfet 1 3 2 dl8 grn rn3c 3.9 k 3 12 dl6 red cpu_vid0 cpu_vid1 cpu_vid2 cpu_vid3 cpu_vid4 cpu_vid5 v ccp gnd + ce10 560uf/4v 5mohm 1 2 + ce1 560uf/4v 5mohm 1 2 + ce4 n/l 5mohm 1 2 + ce8 560uf/4v 5mohm 1 2 + ce5 560uf/4v 5mohm 1 2 + ce2 560uf/4v 5mohm 1 2 + ce9 560uf/4v 5mohm 1 2 + ce3 560uf/4v 5mohm 1 2 + ce6 n/l 5mohm 1 2 + ce7 560uf/4v 5mohm 1 2 vccp gnd c3 22uf/6.3v x5 r 1 2 c7 22uf/6.3v x5 r 1 2 c12 22uf/6.3v x5 r 1 2 c17 22uf/6.3v x5 r 1 2 c1 22uf/6.3v x5 r 1 2 c13 22uf/6.3v x5 r 1 2 c8 22uf/6.3v x5 r 1 2 c4 22uf/6.3v x5 r 1 2 c18 22uf/6.3v x5 r 1 2 c9 22uf/6.3v x5 r 1 2 c14 22uf/6.3v x5 r 1 2 c5 22uf/6.3v x5 r 1 2 c10 22uf/6.3v x5 r 1 2 c15 22uf/6.3v x5 r 1 2 c6 22uf/6.3v x5 r 1 2 c2 22uf/6.3v x5 r 1 2 c11 22uf/6.3v x5 r 1 2 c16 22uf/6.3v x5 r 1 2 06022-019 no t e : ther mi st or ( r th ) used onl y f o r dcr me t hod. do not i nser t for r sense met hod. gr ound pw m 4 f o r 3- phase oper at i on figure 19. typical applications schematic part 1
ADP3194 rev. 0 | page 28 of 32 remove r29 for 3-phase ope ration phase2 pwm1 od# phase4 pwm3 pwm2 od# od# pwm4 cs_ph4 phase3 cs_ph3 cs_ph2 od# phase1 cs_ph1 vcc2 12vr phase_1_rdy phase_2_rdy phase_3_rdy phase_4_rdy gnd vccp vccp vccp gnd gnd +1 2v vccp csref vccp gnd gnd c34 4.7uf/16v x5r(1206) 1 2 jp4 2x2 hdr 1 2 3 4 tp32 rs5 1 tp24 vsws_1 1 r_ph2 10 1 2 sw2 r27 0 ohm c50 4.7uf/16v 1 2 u2 ip2003a 1 2 3 4 5 7 6 8 9 10 vdd enable pwm prdy pgnd pgnd vsw vin vsws1 vsws2 tp34 +5v 1 c33 1uf/16v x5r(0805) 1 2 r_ph4 10 1 2 + ce11 2700uf/16v 12.5x30mm 1 2 c38 4.7uf/16v x5r(1206) 1 2 c37 1uf/16v x5r(0805) 1 2 tp33 rs5 1 c30 4.7uf/16v x5r(1206) 1 2 l5 280nh 30a/0.8mohm 2 1 3 rsense 2 1m ohm 1% c29 1uf/16v x5r(0805) 1 2 rsense5 10 mohm 1% tp14 sw1 1 tp27 phase_2_rdy 1 rsense 1 1m ohm 1% u3 ip2003a 1 2 3 4 5 7 6 8 9 10 vdd enable pwm prdy pgnd pgnd vsw vin vsws1 vsws2 3 2 1 q24 ntd100n02 + ce12 2700uf/16v 12.5x30mm 1 2 jp2 header 2 1 2 1 2 rsense 3 1m ohm 1% pt1 vout 1 23 4 c53 4.7uf/16v 1 2 tp28 vsws_3 1 pt2 gnd out 1 23 4 tp25 phase_1_rdy 1 tp22 gnd 1 tp30 vsws_4 1 r36 10k 1 2 c42 4.7uf/16v x5r(1206) 1 2 c51 4.7uf/16v 1 2 r26 0 ohm tp29 phase_3_rdy 1 tp21 12vr 1 c52 4.7uf/16v 1 2 rsense 4 1m ohm 1% r_ph1 10 1 2 u4 ip2003a 1 2 3 4 5 7 6 8 9 10 vdd enable pwm prdy pgnd pgnd vsw vin vsws1 vsws2 r28 0 ohm u7 ip2003a 1 2 3 4 5 7 6 8 9 10 vdd enable pwm prdy pgnd pgnd vsw vin vsws1 vsws2 l1 0.37uh 18a 2 1 c41 1uf/16v x5r(0805) 1 2 tp17 sw4 1 tp35 -5v 1 tp31 phase_4_rdy 1 l2 280nh 30a/0.8mohm 2 1 3 tp26 vsws_2 1 l3 280nh 30a/0.8mohm 2 1 3 r29 0 ohm + ce15 47uf/16v 5x11mm 1 2 r_ph3 10 1 2 l4 280nh 30a/0.8mohm 2 1 3 tp15 sw2 1 tp16 sw3 1 06022-020 figure 20. typical applications schematic part 2
ADP3194 rev. 0 | page 29 of 32 outline dimensions compliant to jedec standards mo-153-ae 28 15 14 1 8 0 seating plane c oplanarit y 0.10 1.20 max 6.40 bsc 0.65 bsc pin 1 0.30 0.19 0.20 0.09 4.50 4.40 4.30 0.75 0.60 0.45 9.80 9.70 9.60 0.15 0.05 figure 21. 28-lead thin shrink small outline package [tssop] (ru-28) dimensions shown in millimeters ordering guide model temperature range package descript ion package option ordering quantity ADP3194jruz-rl 1 0c to +85c 28-lead tssop 13 reel ru-28 2500 1 z = pb-free part.
ADP3194 rev. 0 | page 30 of 32 notes
ADP3194 rev. 0 | page 31 of 32 notes
ADP3194 rev. 0 | page 32 of 32 notes ?2006 analog devices, inc. all rights reserved. trademarks and registered trademarks are the property of their respective owners. d06022-0-10/06(0)

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